(1) Field of the Invention
The present invention generally relates to a digital subscriber loop interface unit, and more particularly to a digital subscriber loop interface transmitting and receiving information represented by the 2B1Q codes.
(2) Description of the Related Art
In a case where a digital subscriber loop interface unit transmits digital signals represented by the 2B1Q codes using a twisted pair cable like an existing telephone subscriber loop, each transmission path code symbol is one of four values +3, +1, -1, and -3. In addition, for example, a baud rate of 80 kHz is selected as the transmission rate. In such a digital subscriber loop interface unit using the 2B1Q codes, it is required to effectively carry out an echo canceling process and an equalizing process for the input signals.
A conventional digital subscriber loop interface unit using the 2B1Q codes are formed, for example, as shown in FIG. 1. Referring to FIG. 1, the digital subscriber loop interface unit has a hybrid circuit (HYB) 102 connected to a subscriber loop 101, an analog-to-digital converter (A/D) 103, a high pass filter (HPF) 104, an AGC amplifier (AGC) 105, a low pass filter (LPF) 106, a decision feedback equalizer 109. The decision feedback equalizer 109 is formed of a discriminator (DEC) 107 and a transversal equalizer (EQL) 108. The digital subscriber also has a driver (DV) 110, an echo canceler 111 and a digital-to-analog converter (D/A) 115. The echo canceler 111 has a linear echo canceler (LEC) 112, a non-linear echo canceler (NEC) 113 and a jitter echo canceler (JEC) 114.
A transmission signal S(n) is converted, by the digital-to-analog converter 115, into an analog signal having an amplitude which is one of values .+-.3 and .+-.1. The analog signal is supplied from the driver 110 to the subscriber loop 101 via the hybrid circuit 102. A received signal from the subscriber loop 101 is supplied to the analog-to-digital converter 103 via the hybrid circuit 102 so as to be converted into a digital signal in accordance with, for example, an over-sampling analog-to-digital converting technique in the analog-to-digital converter 103. The wave-shaping and the elimination of low-frequency components are then carried out by the high pass filter 104.
Echo components of the received signal is removed from the received signal by the echo canceler 111. The received signal is amplified by the AGC amplifier 105 so as to have a predetermined level and is supplied to the decision feedback equalizer (DFE) 109 via the low pass filter 106. The decision feedback equalizer (DFE) 102 removes inter-symbol interference from the received signal so that data is reproduced.
In a case where the AMI (Alternate Mark Inversion) codes which do not includes DC components are used as transmission codes, even if the hybrid circuit 102 is formed of a hybrid transformer cutting off the DC components, wave form distortion does not occur. On the other hand, in a case where the 2B1Q codes including DC components, if the DC component is cut of the by the hybrid transformer of the hybrid circuit 102, the wave form distortion occurs. As a result, an impulse response shows a characteristic having a long tail. Thus, it is required to increase the number of taps of each of transversal filters in the echo canceler 111 and the decision feedback equalizer 109. In the transversal filter having a large number of taps, the size thereof is large and a large amount of processing is needed so that the dissipation power is increased.
To reduce a degree of the long tail in the impulse response, it is thinkable that the inductance of the hybrid transformer is increased so that the attenuation of the low frequency components is as smaller as possible. However, the hybrid transformer is made larger, so that problems of the cost and volume occur.
Thus, as shown in FIG. 2, it has been known that a primary recursive filter (an IIR filter (Infinite Impulse Response filter)) 121 is added to the transversal filter. Referring to FIG. 2, the transversal filter has a plurality of delay elements (D), multipliers (.times.) and an adder (.SIGMA.) 120. Each of the delay elements (D) delays an input signal by a delay time corresponding to the baud rate. The primary recursive filter 121 has a adder (+), a multiplier (.times.) and a delay element (D). At respective taps each of which is between adjacent delay elements (D), symbols P(n)-P(n-M-1) sampled at sampling times n-(n-M-1) are obtained. The respective taps are connected to the multipliers (.times.), so that the multipliers (.times.) respectively multiplies the symbols P(n)-P(n-M) by tap coefficients CE.sub.1 -CE.sub.M. Multiplying signals output from the multipliers (.times.) are supplied to the adder (.SIGMA.) 120 and added to each other. The symbol P(n-M-1) sampled at the sampling time (n-M-1) and a tap coefficient CE.sub.M+1 are multiplied by the multiplier (.times.), and the multiplying signal is supplied to the primary recursive filter 121. An output signal of the primary recursive filter 121 is added to an adding output signal from the adder (.SIGMA.) 120 by an adder (+). A adding result at the adder (+) is obtained as an equalizing output signal y(n). The tap coefficient CE.sub.M+1 and an attenuation factor are supplied, as parameters, to the primary recursive filter 121. The attenuation factor corresponding to the long tail of the impulse response is equal to or greater than 0.95 and is less than 1.
FIG. 3 shows a unit in which an updating circuit for updating the tap coefficients to be supplied to the transversal filter is added to the circuit shown in FIG. 2. In FIG. 3, those parts which are the same those shown in FIG. 2 are given the same reference numbers. Referring to FIG. 3, an equalizing error signal k.multidot.e(n), obtained by multiplying an error signal e(n) at a sampling time n and a step size k, is supplied to the transversal filter. The symbol P(n) of the received signal at the sampling time n is successively delayed by the delay elements (D) serially connected to each other. The output signals at the respective taps each of which is between adjacent delay elements (D) are multiplied by the equalizing error signal k.multidot.e(n). Each of the multiplying signals is added to a tap coefficient delayed by one period of sampling time, and the adding result is multiplied, one of tap coefficient CE.sub.1 -CE.sub.M+1, by the signal at each tap.
FIG. 4 shows an impulse response of the received signal. In FIG. 4, the axis of ordinate represents an amplitude and the axis of abscissa represents a baud rate time. In addition, a continuous line represents an impulse response in a case where the hybrid transformer has a large inductance, and a dotted line represents an impulse response in a case where the hybrid transformer has a small inductance. Since the hybrid transformer cuts off DC components, the tail of the impulse response generally becomes long. For example, in the case where the hybrid transformer has a large inductance, as indicated by the continuous line, the impulse response proceeds to an attenuation characteristic from a time corresponding to about eighteen taps. On the other hand, in the case where the hybrid transformer has a small inductance, as indicated by the dotted line, the impulse response proceeds to an attenuation characteristic from a time corresponding to about thirty taps.
Referring to the impulse responses indicated by the continuous line and the dotted line in FIG. 4, in the case where the hybrid transformer has the small inductance, the time from which the impulse response proceeds to the attenuation characteristic is delayed in comparison with in the case where the hybrid transformer has the large inductance. As a result, the tail of the impulse response in the case where the hybrid transformer has the small inductance is longer than the tail of the impulse response in the case where the hybrid transformer has the large inductance. In addition, the primary recursive filter 121 acts on a region in which the impulse response is exponentially attenuated. Thus, in the case the hybrid transformer having the small inductance is used, the number of taps located previous to the primary recursive filter 121 must be greater than the number of taps in the case where the hybrid transformer having the large inductance is used.
FIG. 5 shows essential parts of a conventional decision feedback equalizer. Referring to FIG. 5, the decision feedback equalizer has a plurality of delay elements (D), an adder (.SIGMA.) 130, a discriminator 131, adders (+) and multipliers (.times.). Each of the delay elements (D) delays an input signal by a delay time corresponding to the baud rate. When a received signal x(n) is supplied to the decision feedback equalizer, the decision feedback equalizer outputs a received symbol P(n). An equalizing error signal e(n) is generated and the a feedback error signal k.multidot.e(n) is made by using the equalizing error signal e(n).
Received symbols P(n-1)-P(n-N) are respectively multiplied by tap coefficients Cd.sub.1 -Cd.sub.N, and the multiplying signals are added to each other by the adder 130. The adding result is output as an equaling signal y(n) from the adder 130. The equalizing signal y(n) is subtracted from the received signal x(n) so that the received signal x(n) is equalized. The equalized signal is supplied to the discriminator 131, and the discriminating result obtained by the discriminator 131 is used as the received symbol P(n). The decision feedback equalizer is applicable to various types of transmission apparatuses other than the digital subscriber loop interface unit.
In the decision feedback equalizer, a symbol value (A) is multiplied by a tap coefficient (B), and a number (C) is added to the multiplying result. In this case, one of values .+-.3 and .+-.1 is multiplied, as the symbol value (A), by the tap coefficient. Thus, the above calculating process can be carried out by using a 3-inputs adder and a shifter. In addition, it has been known that, in processes in a high pass filter and a low pass filter, a received signal having a plurality of bits and a filter coefficient are multiplied by each other using a shifting process in which the filter coefficient represented by an exponent of "2" is shifted.
In the decision feedback equalizer and the echo canceler in the digital subscriber loop interface unit, the total number of taps is fifty or more. A multiplying process and an adding process must be performed for each tap. Thus, the large number of processes is needed. To simplify processes, the following technique has been proposed (e.g. U.S. Pat. No. 4,926,472).
"+1" is added to each of four kinds of symbol values .+-.3 and .+-.1 of the 2B1Q code so that new four kinds of symbol values 4, 2, 0 and -2 (2.sup.2, 2.sup.1, 0, -2.sup.1) are obtained. According to a set of the new four kinds of symbol values, a process for each tap can be performed by using a 2-inputs adder and a shifter. A set of these new symbol values are referred to, for example, as a "symbol value +1".
Essential parts of a conventional decision feedback equalizer using the "symbol value +1" are shown in FIG. 6. In FIG. 6, those parts which are the same as those shown in FIG. 5 are given the same reference numbers. Referring to FIG. 6, "1" is added to a received symbol P(n) which is discriminated by the discriminator 131 so that the symbol values .+-.3 and .+-.1 of the 2B1Q code is converted into symbol values 4, 2, 0 and -2. A DC (direct-current) correction part 132 corrects a difference between results of operations using the symbol values 4, 2, 0, and -2 obtained by adding +1 to the original symbol values .+-.3 and .+-.1 of the 2B1Q code and results of operations using the original symbol values of the 2B1Q, the operations including a convolution operation and an updating operation of the tap coefficients.
When a j-th tap coefficient, an equalizing error signal and a symbol value at a sampling time n are respectively represented by Cd.sub.j (n), e(n) and P(n), the j-th tap coefficient Cd.sub.j (n+1) at a sampling time (n+1) is represented as follows. EQU Cd.sub.j (n+1)=Cd.sub.j (n)+k.multidot.e(n).multidot.[P(n-j)+1](1)
In addition, the relationship between a tap coefficient Dd(n) at the sampling time n and a tap coefficient Dd(n+1) at the sampling time (n+1), in the DC correction part 132, is represented as follows. EQU Dd(n+1)=Dd(n)+k.multidot.e(n) (2)
FIG. 7 shows a conventional jitter echo canceler in a case where the high pass filter has three taps. Referring to FIG. 7, the jitter echo canceler has an adder (.SIGMA.) 140, switches 141.sub.1 -141.sub.M, delay elements (D) and multipliers (.times.). The symbol P(n) and tap coefficients CJ.sub.11 -CJ.sub.M3 are supplied to this jitter echo canceler.
When insertion or deletion of pulses is performed to obtain a phase locked state in a DPLL (digital phase lock loop), received sampling point is varied. After this, an echo wave form is distorted for a while. To deal with the distortion of the echo wave form, the jitter echo canceler is provided in the digital subscriber loop interface unit. After jitter occurs to cancel the distortion of the echo wave form caused by the variation of sampling points, the convolution operation is performed using all the taps. After this, the switches 141.sub.1 -141.sub.M are controlled so that the number of taps applied to the convolution operation is decreased one by one. That is, the echo of a transmission code output before the jitter occurs is returned after the jitter has occurred (after the sampling time is changed), and the value of the echo differs from the value of the echo obtained in a case of no jitter since the sampling time is changed. To correct this difference, the jitter echo canceler is operated in addition to the linear echo canceler and the non-linear echo canceler.
Then, after the jitter has occurred, the transmission signal is transmitted in synchronism with new sampling times, so that the echo of the transmission signal output after the jitter has occurred is corrected by only the linear echo canceler and the non-linear echo canceler.
The output of the jitter echo canceler is a function of the transmission code S(n), and the impulse response is at an approximately constant value after I periods have elapsed from a time at which the jitter has occurred. The output of the jitter echo canceler is obtained by carrying out the convolution operation until the variation of the amplitude cased by changing the sampling time is out for M periods. That is, the output y(I) obtained after the I periods had elapsed from the time at which the jitter has occurred is represented as follows. EQU y(I)=[.SIGMA..sub.j=I.sup.M {CJ.sub.Ij .multidot.S(I-j)}].multidot.jdr(3)
In the above equation, .SIGMA..sub.j =I.sup.M means an adding operation from j=I to j=M, that is, indicates the convolution operation. The number I of periods falls within a rage 1-M (1, 2, . . . , and M), and "jdr" represents a direction of the jitter and has 1 or-1. The value of the "jdr" is determined as 1 or -1 based on whether, in the DPLL circuit (not shown in figures), the pulse insertion is performed to advance the sampling phase or the pulse deletion is performed to delay the sampling phase. S(I-j) represents the transmission code as a function of the number I of periods which have elapsed, and S(1) represents the transmission code immediately after the jitter has occurred.
The tap coefficient CJ.sub.Ij is represented as a function of the number I of periods, because the high pass filter formed of the transversal filter is provided at a front end of the jitter echo canceler. For example, in a case where the jitter occurs at a time n, the echo related to the transmission code previous to the P(n-1) is affected by changing the sampling time. However, even if the echo is related to the transmission code previous to P(n-1), signals which have been already stored in the delay elements in the high pass filter are not affected. Thus, until the signals are output from the high pass filter, the tap coefficients of the jitter echo canceler are not constant, and after this, the tap coefficients becomes constant. Thus, when representing the number of taps of the transversal filter forming the high pass filter as K, the tap coefficient CJ.sub.Ij is constant in a rage of I.gtoreq.K.
In an example of the conventional jitter echo canceler shown in FIG. 7, the number K of the taps of the high pass filter is set at 3. The switches 141.sub.1 -141.sub.M are switched as shown in FIG. 7 at a time corresponding to the period I=1. Thus, the tap coefficients CJ.sub.11 -Cj.sub.M1 are supplied to this jitter echo canceler. In the next period I=2, the tap coefficients CJ.sub.22 -CJ.sub.M2 are supplied thereto. In the period I=3, the tap coefficients CJ.sub.33 -CJ.sub.M3 are supplied thereto, and in the period I=4, the tap coefficients CJ.sub.43 -CJ.sub.M3 are supplied to this jitter echo canceler.
The updating equation, corresponding to the equation (3), for the tap coefficient is represented as follows. EQU CJ.sub.Ij (I+1)=CJ.sub.Ij (I)+k.multidot.e(n).multidot.S(I-j).multidot.jdr(4)
In a rage of I.gtoreq.K, EQU CJ.sub.Ij =CJ.sub.Kj ( 5)
stands, where K is the number of taps of the high pass filter.
In the digital subscriber loop interface unit shown in FIG. 1, the low pass filter 106 at the front end of the decision feedback equalizer 109 forms a wave of post-cursor portion after the main pulse. Hereinafter the low pass filter 106 is referred to as a post-cursor equalizer. The transfer function of this post-cursor equalizer can be represented by EQU 1/(1-a.multidot.z.sup.-). (6)
The filter coefficient a is equal to zero (a=0) when the cable length is minimum, and the filter coefficient a is equal to 0.75 (a=0.75) when the cable length is maximum.
It can be determined, based on the gain of the AGC amplifier 105, whether or not the cable length is large or small. When the cable length is large, the received signal is attenuated. Thus, the AGC amplifier 105 amplifies the received signal with a large gain. On the other hand, when the cable length is small, the received signal is not almost attenuated, so that the AGC amplifier 105 amplifies the received signal with a small gain. Thus, in accordance with the gain of the AGC amplifier 105, the filter coefficient a of the post-cursor equalizer is changed. In a case where the above post-cursor equalizer is used, there is a problem in that a pull-in time (a time required to obtain a signal converging on a stable state) in a case of using a long cable may be greater than that in a case of using a short cable. Because the gain of the AGC amplifier is large in use of the long cable as has been described above, noise components are also enlarged so that a noise error becomes large, the pull-in time may be large.
The problem of the long pull-in time may also depend on the difference between the main pulse forms in use of the long cable and in use of the short cable. the impulse response at the output of the low pass filter 106 is shown in FIG. 18. In the case where the length of the cable is large, the half width of the main pulse is greater than the baud rate period T, as shown in FIG. 18(a). On the other hand, in the case where the length of the cable is small, the half of the main pulse is less than the baud rate period T, as shown in FIG. 18(b). Due to changing the parameter a of the high pass filter 104, the half width of the main pulse can be close to th baud rate period T. However, the parameter of the high pass filter 104 is set at a value so that the whole post-cursor portion of the impulse response, rather than the width of the main pulse, has a pulse amplitude in post-cursor range as small as possible. Thus, when the length of the cable is large as has been described above, the half width of the main pulse in the impulse response is large.
The sampling phase is generally not constant at start of the pull-in operation, so that the optimum sampling phase is searched for by moving the sampling point in a time axis in the phase extracting circuit (not shown in figures). According to an algorithm of the decision feedback equalizer, the main-cursor (the main response) corresponds to an amplitude at a sampling time at which the maximum value is obtained among a plurality of sampling values sampled every baud rate period T. The pre-cursor corresponds to an amplitude at a time previous by T to the sampling time at which the main-cursor is obtained. The sampling phase corresponding to the pre-cursor is normally determined at zero. The post-cursor corresponds to an amplitude at each of times delaying from the main-cursor by nT where n represents positive integers. For example, in FIG. 8(a) and (b), the pre-cursor is zero and sampling points a1 and b1 delaying from the precursor by the period T is the optimum phases for the pull-in process.
The sampling points a1 and a3 are separated by the period T, and the amplitudes at the sampling points a1 and a3 are equal to each other. The sampling points b1 and b3 has the same relationships as the sampling points a1 and a3. In this case, if the pull-in process starts from a sampling phase which is prior to the optimum sampling points a1 and b1 and after the sampling points a2 and b2, a value at a sampling point further previous by the period T to each of the sampling points a2 and b2 is negative. Thus, a phase control is carried out so that the sampling phase is delayed. On the other hand, if the pull-in process starts from a sampling phase which is after the optimum sampling points a1 and b1 and prior to the sampling points a3 and b3, a value at a sampling point previous by the period T to each of the sampling points a3 and b3 is positive. Thus, a phase control is carried out so that the sampling phase advances. Thus, in both the case, the phase control is carried out so that the sampling phase is brought close to each of the optimum sampling points a1 and b1. However, in a case where the main pulse is wide, the difference between the amplitude at each of the sampling points a2 and a3 and the amplitude at the optimum sampling point a1 is small, so that both the amplitudes are large in a wide range. As a result, it is not clear which amplitude corresponds to the main response. Thus, the pull-in time is increased.
In a case shown in FIG. 8(b), the amplitude at each of the sampling points b2 and b3 is less than the amplitude at the optimum sampling point b1. If sampling points are slightly moved from the phases including the sampling points b2 and b3, the amplitude at one sampling point is much larger than the amplitude at another sampling point. As a result, it is clear which sampling point corresponds to the main pulse. Thus, the pull-in time can be short.
In concrete terms, in the case where the length of the cable is large, the frequency at which decision errors occur in initial step of the pull-in process in the decision feedback equalizer is lager than the frequency at which the decision errors occur in the case where the length of the cable is short. Thus, a long time is required to obtain right tap coefficients in the coefficient updating operation represented by the equation (1).
In the conventional digital subscriber loop interface unit formed as shown in FIG. 1, the 2B1Q code is used for the transmission of information. Since the hybrid circuit 102 is formed of the hybrid transformer, the transmission characteristic depends on the impedance of each of the hybrid transformers in both the transmitting side unit and the receiving side unit. Thus, even if the recursive filter is used, the number of taps of each of the echo canceler 111 and the decision feedback equalizer 109 must be twenty four or more. That is, the recursive filter is used only as aid.
In a subscriber side of the digital subscriber loop interface unit, after the pull-in process in the echo canceler 111 is completed by using the transmission signal to be transmitted to a destination, the taps of the echo canceler 111 are fixed. The pull-in process in the decision feedback equalizer 109 is then performed using the received signal. In the switching unit side, the transmission timing is fixed, so that the sampling phase must be synchronized with the timing of the received signal. Thus, the sneak time of the echo can not be fixed. As a result, in the pull-in process, the number of taps of both the echo canceler 111 and the decision feedback equalizer must be simultaneously updated.
In this case, if the tap coefficients are updated using the "symbol value +1" as the symbol in both the echo canceler 111 and the decision feedback equalizer 109, there is a case where the pull-in is incomplete. This is caused by using the same error in both the echo canceler 111 and the decision feedback equalizer 109. In general, each tap coefficient is calculated by based on the correlation between the symbol P(n-j) +1 and the error e(n) as indicated by the equation (1). In the echo canceler 111, the tap coefficients are obtained by using the transmission symbol, and in the decision feedback equalizer 109, the tap coefficients are obtained by using symbols reproduced from the received signals. That is, the tap coefficients are separately obtained in the echo canceler 111 and the decision feedback equalizer 109.
However, the value of the DC correction part indicated by the equation (2) depends on only the error e(n). When the pull-in processes in the echo canceler 111 and the decision feedback equalizer 109 are simultaneously performed using the same error e(n), the respective DC correction parts performs the same process. As a result, in the respective the echo canceler 111 and the decision feedback equalizer 109, the correction of DC components required by them are not always performed.
In addition, the echo canceler 111 has the non-linear echo canceler 113 which functions as the DC correction part. As a result, the echo canceler 111 may not be provided with the DC correction part. On the other hand, in the decision feedback equalizer 109, since the DC component are not ignored, the decision feedback equalizer 109 must be provided with the DC correction part for correcting the DC components.
As has been described above, in a case where both the echo canceler 111 and the decision feedback equalizer 109 carries out the convolution operation and the updating operation of the tap coefficient using the "symbol value +1", both the echo canceler 111 and the decision feedback equalizer 109 are in a state where there is substantially the DC correction part. Thus, in an initial state and/or in an step size, the tap coefficients are updated so that the non-linear echo canceler 113 corrects DC components which should be corrected by the DC correction part of the decision feedback equalizer 109. As a result, the tap coefficients reach limited value, and the non-linear echo canceler 113 can not perform the correction of non-linear components to be processed. In this case, the pull-in may be incomplete.
The tap coefficients of the transversal transformer of the decision feedback equalizer 109 correspond to the post-cursor portion of the impulse response, but the transversal transformer has no tap coefficients corresponding to the pre-cursor and the main-cursor. If the value of the DC correction part is limited to an item corresponding to the post-cursor (the value of the DC correction part is the sum of coefficients of only the post-cursor portion, and is close to "1" not close to zero.), the calculation of [(the value of the main-cursor).times.(the symbol value P(n)] must be performed when the error is calculated. As a result, the number of calculations can not be decreased.
If the term corresponding to the main-cursor is considered as the DC correction part, the value of the DC correction part is close to zero. In this case, the error e(n) is obtained in accordance with a calculation [(the main-cursor value Cd.sub.0).times.(the new symbol P(n)+1)]. As a result, the number of calculation is not increased. However, in this case, the control of the pull-in is complex. For example, before the pull-in of the tap coefficients in the decision feedback equalizer 109 is performed, the tap coefficient Cd.sub.0 corresponding to the main-cursor is calculated in a state where the tap coefficient corresponding to the post-cursor in the decision feedback equalizer 109 is maintained at zero. The AGC gain is then determined so that the calculated tap coefficient Cd.sub.0 falls within a predetermined range. If this AGC process is performed, since the coefficient value Cd.sub.0 corresponding to the main-cursor is not zero, the DC correction part can not be zero. The value of the DC correction part obtained when the AGC process is completed is approximately the same as the value of the main-cursor so as to be greatly different from zero. After this, the pull-in of the tap coefficient in the decision feedback equalizer 109 is performed. When the tap coefficient corresponding to the post-cursor is close to the final value, the DC correction part is gradually varied to a value close to zero.
As has been described above, the DC correction part in the decision feedback equalizer 109 has a large value once in the pull-in process, and is then varied so as to return to zero. Since the coefficient varies complexly as this, the step size must be filly controlled. As a result, the control is complex and the time required for the pull-in process is long.
Although the linear echo canceler 112 has about thirty taps, the jitter echo canceler 114 forming a part of the echo canceler 111 has the number of taps falling within 7-9. The convolution range is varied with elapsing time, so that the tap coefficients are not constant. Thus, even if the "symbol value +1" is applied to the jitter echo canceler 114, the DC correction part which is unchangeable with time can not respond to the "symbol value +1". That is, the "symbol value +1" can not be applied to the conventional jitter echo canceler 114.
The characteristics of the post-cursor equalizer (the low-pass filter 106) is selected so that the post-cursor portion of the impulse response, that is, the tail of the impulse response has a predetermined shape. Thus, if the cable is long, the emphasis of the high frequency components is insufficient so that the main pulse expands. As a result, the probability that an error occurs in the decision feedback equalizer 109 immediately after the pull-in process starts is increased, and the time required for the pull-in process is long.